This research presents a triple-aperture waveguide antenna as the primary feed of parabolic reflectors. The proposed antenna is able to rectify the asymmetry and also achieve a symmetrical unidirectional beam through the application of two parasitic coupling apertures. The design of the antenna is that of a rectangular waveguide (radiating aperture) vertically jointed to the two coupling apertures of the same measurement widthwise (i.e., one stacked on top and the other underneath) to achieve the symmetrical beam. The rectangular waveguide is 97.60 mm and 46.80 mm in width (a) and height (b), respectively, to propagate the WLAN frequency band of 2.412–2.484 GHz. Simulations were carried out to determine the optimal antenna parameters and an antenna prototype was subsequently fabricated and tested. The simulated beamwidths in the E- and H-planes at -3 dB were equally 67° (i.e., 67° for both the E- and H-planes) and at -10 dB also equally 137°, while the measured results at -3 dB were equally 65° and at -10 dB equally 135°. The simulation and measured results are thus in good agreement. The simulated and measured antenna gains are, respectively, 8.25 dBi and 9.17 dBi. The findings validate the applicability of the antenna as the prime feed for rotationally symmetric parabolic reflectors.
1. Introduction
In recent decades, the point-to-point communications systems have rapidly advanced and become one of the brightest areas of the communications business [1]. Specifically, the point-to-point links between hosts and clients are required in several wireless systems for communication over long distances, such as the microwave radio relay link, long length Wi-Fi, wireless WAN/LAN link, satellite communication, and home satellite television [2].
Horn antennas, which are a principal component of the point-to-point communication systems, were first developed nearly a century ago for military and scientific purposes but were not widely adopted until World War II. Typical horn antennas are of either rectangular or conical structures. The rectangular structure can further be divided into the H-plane, E-plane, and pyramidal horn structures [3, 4].
To enhance the performance of the point-to-point communication requires a narrow-beam antenna with high gain [5], and one possible method to achieve the narrow beam is through a parabolic metal reflector antenna. In addition, the feeding point of the parabolic reflector should be located at the reflector focus to generate the narrow beam (i.e., pencil beam). In [6], the authors reviewed publications on the point-to-point communication and documented that the generation of the pencil beam requires an antenna with symmetrical beam as the primary feed.
The radiation pattern of a parabolic reflector (secondary antenna) typically corresponds to that of the primary feed antenna. In other words, the asymmetric radiation pattern from the primary feed contributes to the asymmetrical incidence of the secondary antenna and vice versa [7]. In fact, it is difficult to obtain the symmetrical radiation pattern at the primary feed due to the beamwidth asymmetry between the E- and H-planes [8].
To address the asymmetry issue, several antenna structures have been proposed. In [9], a pyramidal horn antenna was utilized to obtain the symmetric radiation patterns in the E- and H-planes; nevertheless, the antenna structure was large with the width, height, and length of 2.80λ, 2.03λ, and 4.35λ. In [10], the author experimented with a diagonal horn antenna with very low cross-polarization and axially symmetrical field distribution at the horn aperture; however, the antenna has a large aperture size of 5.00λ and 5.00λ in width and height.
Generally, the corrugated horn consisting of parallel slots or grooves can create a hybrid-mode pattern in the aperture which straightens out the electric field and reduces diffractions from the edge [11, 12]. In [13–18], the conical corrugated horn antennas were utilized to generate the symmetrical beam with high gain and low side lobe level. The antennas structures are however complicated. On the contrary, for the typical rectangular corrugated horn antenna structure, the slots or grooves are located inside both vertical and horizontal planes around the rectangular aperture area. The rectangular corrugated horn [19] was experimented and it was reported that this antenna type failed to generate the symmetrical beam despite its relatively large size. In [20], the authors proposed a flared rectangular horn corrugated along the E-plane flaring walls to achieve the symmetry in the E- and H-planes; nevertheless, the antenna construction is very complicated and the antenna aperture is considerably large. For the rectangular waveguide aperture perpendicular to the z direction, the diffracted wave is almost in yz-plane and it is minimal in xz-plane. The parasitic coupling apertures are proposed to locate in yz-plane (vertical plane) outside the radiating aperture. One aperture is stacked on top (top section) and the other is stacked underneath (bottom section) the radiating rectangular waveguide (middle section). This makes the proposed triple-aperture waveguide antenna less complicated and easy fabricated compared to the typical corrugated horn antenna.
This research has thus proposed a triple-aperture waveguide antenna as the primary feed of a parabolic reflector. The antenna can achieve a symmetrical unidirectional beam through the use of two parasitic coupling apertures. In fact, an independent use of an open-ended rectangular waveguide as the primary feed of a parabolic reflector is rare due to the asymmetry and low directivity [21]. The proposed antenna design consists of a rectangular radiating waveguide jointed to the two coupling apertures, one stacked on top and the other underneath. In this research, the rectangular waveguide is 97.60 mm and 46.80 mm in width (a) and height (b), respectively, to propagate the WLAN frequency band of 2.412–2.484 GHz. The proposed antenna is for point-to-point communication and thus requires a front feed parabolic reflector [5, 22, 23] and is appropriate for WLAN applications along IEEE 802.11b/g/n. Moreover, the technique of coupling apertures can be applied to other frequency ranges by adjusting the antenna electrical size.
The organization of the research is as follows: Section 1 is the introduction. Section 2 describes the design of the proposed triple-aperture waveguide antenna, while Section 3 discusses the parametric study and the simulation results of the antenna. Section 4 deals with the antenna prototype and the experimental results. The concluding remarks are provided in Section 5.
2. The Antenna Design
The structure of the triple-aperture waveguide antenna, as the name implies, is made up of one radiating section and two coupling sections. The configuration of the proposed antenna is illustrated in Figure 1, in which the radiating aperture is the middle section with the width and height of a and b, while the two coupling apertures refer to those on top and underneath the radiating aperture, with the width, height, and length of a, b1, and l2. The width (a=97.60 mm) and height (b=46.80 mm) of the rectangular waveguide (the middle section) are fixed, where the relationship between its width (a) and length (b) is that of a=2b, to achieve the center frequency of 2.45 GHz in the dominant mode (TE10) [24–28].
Geometry of the proposed antenna.
Inside the middle section (the radiating section) is mounted a linear electric probe at a distance of 0.25λ from its closed end (l1) along the y direction with the axial z-axis propagation. The height of linear electric probe (l) is 0.25λ. The inclusion of both parasitic coupling apertures on the vertical plane at the open end is to reduce the diffraction field and achieve symmetry.
In this research, the initial length of the radiating section (L) is 0.75λg because the standing wave pattern is repeated every 0.50λg. Therefore, the distance between the maximum and minimum electric field distributions is 0.25λg and the entire length of the radiating section (L) is approximately 0.75λg so as to realize the maximum electric field density at the aperture, as shown in Figure 2. The length of the rectangular radiating waveguide (L) is thus 152.40 mm [29].
The simulated electric field distribution of the radiating section of the proposed antenna.
3. Parametric Study and Simulation Results3.1. The Antenna without Parasitic Coupling Apertures
Under this scenario, simulations were carried out on a waveguide without the parasitic coupling apertures using CST Microwave Studio. The waveguide is of rectangular shape, where the relationship between its width (a) and length (b) is that of a=2b. In addition, the waveguide is of aluminum material and 2 mm in thickness. Inside the waveguide is an electric probe that is connected to a 50 Ω N-type connector. The width (a) and length (b) of the rectangular waveguide are capable of achieving the resonant frequency (fr) at the center frequency of 2.45 GHz. Table 1 tabulates the physical and electrical sizes of the rectangular waveguide (i.e., the antenna without the coupling apertures).
Parameters of the antenna without the parasitic coupling apertures.
Parameters
Physical size (mm)
Electrical size
a
97.60
0.797λ
b
46.80
0.382λ
L
152.40
1.244λ
l
30.00
0.244λ
l1
52.40
0.427λ
Figure 3 illustrates the simulated electric field distribution at the center of the width (a) side of the antenna without the parasitic coupling apertures at the 2.45 GHz frequency. It is found that the distribution travels in the z direction and that the diffraction grows stronger around the edges of the waveguide (the yz-plane). On the other hand, the diffraction on the xz-plane is minimal. The yz-plane phenomenon contributes to the asymmetry between the E- and H-plane radiations.
The simulated electric field distributions at the center of the width (a) side of the antenna without parasitic coupling apertures at 2.45 GHz (side view).
Figures 4 and 5, respectively, depict the simulated S11 and gain as well as the input impedance of the antenna without the parasitic coupling apertures. As illustrated in the figures, in the absence of the coupling apertures, the antenna gain (6.70 dBi) is relatively low despite the fairly satisfactory input impedance (Z0=48.75+j4.44 Ω) and S11<-10 dB.
Simulated S11 and gains relative to frequency of the antenna without parasitic coupling apertures.
Input impedance of the antenna without parasitic coupling apertures.
In Figure 6, the simulated beamwidths at -3 dB and -10 dB in the E-plane, respectively, are 106° and 270° and in the H-plane are 65° and 110°. Since the principle application of the proposed triple-aperture waveguide antenna is the primary feed of a parabolic reflector, this research has thus taken into account the beamwidth at -10 dB. As illustrated in Figure 6, the radiation patterns of the antenna without the parasitic coupling apertures in the E- and H-planes are asymmetrical.
The simulated radiation patterns of the antenna without parasitic coupling apertures at 2.45 GHz: (a) E-plane and (b) H-plane.
3.2. The Antenna with Parasitic Coupling Apertures
To rectify the asymmetry, two parasitic coupling apertures are incorporated with the rectangular waveguide whereby one aperture is stacked on top (top section) and the other underneath (bottom section) the radiating waveguide (middle section). To achieve the resonant frequency at the center frequency of 2.45 GHz, this research has utilized the TE_{101} mode to determine the length (l2) and height (b1) of the parasitic coupling apertures [28, 29] while its width (a) is identical to that of the radiating rectangular waveguide (97.60 mm). Simulations were then carried out using CST Microwave Studio and the optimal simulation results for the waveguide antenna with the parasitic coupling apertures are tabulated in Table 2. The realization of the target resonant frequency is largely subject to a and b1.
Parameters of the antenna with the parasitic coupling apertures.
Parameters
Physical size (mm)
Electrical size
a
97.60
0.797λ
b
46.80
0.382λ
L
152.40
1.244λ
l
30.00
0.244λ
l1
52.40
0.427λ
b1
28.98
0.236λ
l2
50.80
0.414λ
Figure 7 illustrates the simulated electric field distribution of the waveguide antenna with the parasitic coupling apertures at the center of the width (a) side. It is found that the wave diffraction is reduced with the use of the coupling apertures (the top and bottom shorter sections) in conjunction with the radiating waveguide (the middle section).
The simulated electric field distributions at the center of the width (a) side of the antenna with parasitic coupling apertures at 2.45 GHz (side view).
3.2.1. Impedance Bandwidth for Various Antenna Parameters
Figure 8 depicts the simulated S11<-10 dB for various rectangular waveguide lengths (L) and at L=1.244λ the simulated S11 resonates at the center frequency of 2.45 GHz. Figure 9 illustrates the simulated S11<-10 dB for various electric probe heights (l) and the results indicate the optimal height of the probe (l) of 0.245λ. In Figure 10, the simulated S11 for various distances between the probe and the closed end of the radiating waveguide (l1) reveal that the optimal distance that achieves the resonance at the target center frequency is 0.414λ.
Simulated S11 for various waveguide lengths (L).
Simulated S11 for various electric probe heights (l).
Simulated S11 for various distances between the probe and the closed end of the waveguide (l1).
Figure 11 illustrates the simulated S11 of the proposed waveguide antenna for various heights (b1) of the coupling aperture, which were varied between 0.057λ and 0.449λ. The results indicate that b1 exerts little influence over S11 and the optimal coupling aperture height (b1) is 0.236λ, at which the beamwidths at -3 dB and -10 dB in both the E- and H-planes are symmetrical. This confirms that the utilization of the coupling apertures contributes to the reduction of the diffraction field at the edges of the radiating waveguide. Figure 12 depicts the simulated S11 for various lengths of the coupling aperture (l2) and the results show that S11 vary considerably with the variation in l2. The resonant frequency is however achieved at l2 of 0.427λ.
Simulated S11 for various heights of the parasitic coupling apertures (b1).
Simulated S11 for various lengths of the parasitic coupling apertures (l2).
3.2.2. Radiation Patterns for Various Heights of the Parasitic Coupling Apertures (<inline-formula>
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Figures 13–15, respectively, illustrate the simulated radiation patterns for various heights of the parasitic coupling apertures (b1) in the E- and H-planes at the lower, center, and upper frequencies of 2.412, 2.45, and 2.484 GHz, where b1 was varied between 0.057λ and 0.449λ. Interestingly, in the H-plane, the beamwidths at -3 dB (HPBW) and -10 dB for the three frequencies exhibit slight differences. On the other hand, those in the E-plane for the aperture heights (b1) between 0.057λ and 0.155λ are noticeably wider than the corresponding beamwidths for the three frequencies in the H-plane. For b1 of 0.236λ and 0.351λ, the beamwidths at both -3 dB and -10 dB in both planes at the three frequencies are symmetrical. Thus, the coupling aperture height of 0.236λ is selected due to the smallest cross-sectional area with the symmetrical radiation pattern. Table 3 tabulates the beamwidths at -3 dB (HPBW) and -10 dB for the three frequencies in the E- and H-planes for the various heights of the parasitic coupling apertures (b1).
Simulated beamwidths at −3 dB and at −10 dB for various heights of the parasitic coupling apertures (b1) in the E- and H-planes for the lower, center, and upper frequencies of WLAN.
Aperture heights (b1)
2.412 GHz
2.45 GHz
2.484 GHz
HPBW
Beamwidth
HPBW
Beamwidth
HPBW
Beamwidth
(−3 dB)
−10 dB
(−3 dB)
−10 dB
(−3 dB)
−10 dB
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
b1=0.057λ
104°
67°
200°
124°
208°
66°
202°
124°
105°
64°
202°
124°
b1=0.155λ
78°
66°
196°
130°
74°
64°
196°
130°
72°
63°
128°
196°
b1=0.236λ
67°
67°
137°
137°
67°
67°
137°
137°
66°
66°
138°
138°
b1=0.351λ
69°
69°
144°
142°
69°
69°
142°
142°
69°
69°
142°
142°
b1=0.449λ
70°
71°
154°
148°
70°
70°
154°
148°
71°
70°
152°
148°
Simulated radiation patterns for various parasitic coupling aperture heights (b1) at the lower frequency of 2.412 GHz in (a) the E-plane and (b) the H-plane.
Simulated radiation patterns for various parasitic coupling aperture heights (b1) at the center frequency of 2.45 GHz in (a) the E-plane and (b) the H-plane.
Simulated radiation patterns for various parasitic coupling aperture heights (b1) at the upper frequency of 2.484 GHz in (a) the E-plane and (b) the H-plane.
3.2.3. Radiation Patterns for Various Lengths of the Parasitic Coupling Apertures (<inline-formula>
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Figures 16–18, respectively, illustrate the simulated radiation patterns for various lengths of the parasitic coupling apertures (l2) in the E- and H-planes at the lower, center, and upper frequencies of 2.412, 2.45, and 2.484 GHz, where l2 was varied between 0.329λ and 0.525λ. It is found that at l2 of 0.427λ the symmetry in both planes for the beamwidths at -3 dB and -10 dB is achieved. Table 4 tabulates the beamwidths at -3 dB (HPBW) and -10 dB for the three frequencies in the E- and H-planes for the various lengths of the parasitic coupling apertures (l2).
Simulated beamwidths at −3 dB and at −10 dB for various lengths of the parasitic coupling apertures (l2) in the E- and H-planes for the lower, center, and upper frequencies of WLAN.
Rectangular aperture lengths (l2)
2.412 GHz
2.45 GHz
2.484 GHz
HPBW
Beamwidth
HPBW
Beamwidth
HPBW
Beamwidth
(−3 dB)
−10 dB
(−3 dB)
−10 dB
(−3 dB)
−10 dB
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
l2=0.329λ
65°
69°
210°
142°
65°
68°
206°
142°
65°
68°
202°
140°
l2=0.378λ
66°
68°
158°
138°
65°
67°
148°
138°
65°
67°
146°
136°
l2=0.427λ
67°
67°
137°
137°
67°
67°
137°
137°
66°
66°
138°
138°
l2=0.476λ
71°
68°
138°
136°
70°
67°
138°
134°
70°
66°
138°
132°
l2=0.525λ
75°
68°
138°
134°
74°
67°
140°
134°
73°
67°
140°
132°
Simulated radiation patterns for various parasitic coupling aperture lengths (l2) at the lower frequency of 2.412 GHz in (a) the E-plane and (b) the H-plane.
Simulated radiation patterns for various parasitic coupling aperture lengths (l2) at the lower frequency of 2.45 GHz in (a) the E-plane and (b) the H-plane.
Simulated radiation patterns for various parasitic coupling aperture lengths (l2) at the lower frequency of 2.484 GHz in (a) the E-plane and (b) the H-plane.
Table 5 compares the simulated -3 dB and -10 dB beamwidths of the proposed antenna with and without the parasitic couplings apertures at the frequencies of 2.412, 2.45, and 2.484 GHz. Without the coupling apertures, the beamwidths at -3 dB and -10 dB in the E-plane and those in the H-plane for the three frequencies are dissimilar; in other words, the incidence of asymmetry is observed. Nevertheless, with the parasitic coupling apertures on top and underneath the width (a) side of the radiating rectangular waveguide, the -3 dB and -10 dB beamwidths in both planes for the three frequencies become symmetrical.
Simulated beamwidths at −3 dB and at −10 dB of the proposed antenna with and without the parasitic coupling apertures at the lower, center, and upper frequencies of WLAN.
Frequency (GHz)
HPBW (−3 dB beamwidth)
−10 dB beamwidth
Without parasitic coupling apertures
With parasitic coupling apertures
Without parasitic coupling apertures
With parasitic coupling apertures
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
E-plane
H-plane
2.412
105°
65°
67°
67°
275°
116°
137°
137°
2.450
106°
64°
67°
67°
275°
116°
137°
137°
2.484
108°
63°
66°
66°
276°
116°
138°
138°
Figure 19 depicts the simulated antenna gains for various parasitic coupling aperture heights (b1). For b1=0.236λand0.449λ, the antenna gains are relatively similar for the entire WLAN frequency range. The aperture height (b1) of 0.236λ is thus selected for the smallest cross-sectional area with symmetrical pattern. Figure 20 illustrates the simulated input impedance of the proposed antenna with the coupling apertures. The input impedance (Z0) at the center frequency of 2.45 GHz is 50.12+j0.77 Ω.
Simulated antenna gains for various heights of the parasitic coupling apertures (b1).
Simulated input impedance of the proposed antenna with the parasitic coupling apertures.
When comparing the proposed triple-aperture waveguide antenna with the conventional pyramidal horn antenna, it is obvious that the proposed antenna possesses smaller total antenna length. The length of the probe-fed waveguide must be appropriately designed to achieve the acceptable impedance matching [30]. The total length of the horn antenna is composed of the length of horn aperture and waveguide feeder. However, the radiating aperture and waveguide feeder of the proposed antenna are integrated into single structure. Therefore, the proposed antenna requires smaller total length to achieve the acceptable impedance matching. For instance, the total length of the conventional pyramidal horn antenna is 18.5% larger than that of the proposed antenna structure with the same radiating aperture and antenna gain. In addition, the proposed triple-aperture waveguide antenna has 11-dB lower cross-polarization level than the conventional pyramidal horn antenna with the same antenna size. The reason is that the larger width of the pyramidal horn causes the higher horizontal electric field component.
4. Experimental Results
Figure 21 presents photographs of the prototype of the proposed triple-aperture waveguide antenna. The proposed antenna is operable in the 2.30–2.60 frequency range (12.245%), which also covers the WLAN frequency, for S11<-10 dB. The antenna prototype was fashioned from aluminum of 2 mm in thickness. The radiation pattern of the prototype antenna is symmetrical with unidirectional pattern, rendering it appropriate for use in the front feeding parabolic reflector in the point-to-point communication.
Photographs of the antenna prototype: (a) perspective view, (b) front view, and (c) side view.
Figure 22 compares the simulated and measured S11 of the waveguide antenna with the parasitic coupling apertures. The simulation and measured results show the operating range of the antenna of 2.28–2.80 GHz and 2.30–2.60 GHz, respectively, indicating their good agreement.
The simulated and measured S11 of the waveguide-dependent triple-aperture antenna.
Figure 23 compares the simulated and measured beamwidths at 2.45 GHz in the E- and H-planes. Meanwhile, Table 6 tabulates the -3 dB and -10 dB beamwidths in both planes for the three frequencies. In Figure 24, the simulation antenna gain in front of the antenna at 0° is compared against the measured gain. The proposed antenna could achieve the maximum gain of 9.17 dBi at the WLAN center frequency.
Measured beamwidths at −3 dB and at −10 dB of the proposed antenna with the parasitic coupling apertures in the E- and H-planes at the lower, center, and upper frequencies of WLAN.
Frequency (GHz)
HPBW (−3 dB beamwidth)
−10 dB beamwidth
E-plane
H-plane
E-plane
H-plane
2.412
65°
65°
135°
135°
2.450
65°
65°
135°
135°
2.484
65°
65°
135°
135°
The simulated and measured radiation patterns at 2.45 GHz: (a) E-plane and (b) H-plane.
The simulated and measured antenna gains in front of the antenna (0°).
5. Conclusion
This research has proposed the triple-aperture waveguide antenna as the primary feed of parabolic reflectors. The antenna could rectify the asymmetry and achieve the symmetrical unidirectional beam with the incorporation of two parasitic coupling apertures. The antenna consists of the radiating rectangular waveguide vertically jointed to the two coupling apertures to achieve the symmetrical beam. The rectangular waveguide is 97.60 mm and 46.80 mm in width (a) and height (b), respectively, to propagate the WLAN frequency band of 2.412–2.484 GHz. In this research, the CST Microwave Studio program was deployed in simulation to determine the optimal antenna parameters and an antenna prototype was subsequently fashioned and experimented. The simulated beamwidths in the E- and H-planes at -3 dB were equally 67° and at -10 dB equally 137°, while the measured beamwidths at -3 dB were equally 65° and at -10 dB equally 135° in the E- and H-planes. The simulation and measured results are in good agreement. The simulated and measured antenna gains are, respectively, 8.25 dBi and 9.17 dBi. The findings confirm the applicability of the antenna as the prime feed for rotationally symmetric parabolic reflectors. In addition, the proposed antenna is light, inexpensive, and of low profile.
Competing Interests
The authors declare that there are no competing interests regarding the publication of this paper.
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