International Journal of Antennas and Propagation
Volume 2019 (2019), Article ID 6463101, 12 pages
https://doi.org/10.1155/2019/6463101

Research Article

A Compact CPW-Fed Low-Profile Wideband Circularly Polarized Slot Antenna with a Planar Ring Reflector for GNSS Applications

Jiade Yuan, Yujie Li, Zhimeng Xu, and Jiamin Zheng

College of Physics and Information Engineering, Fuzhou University, Fuzhou 350116, China

Correspondence should be addressed to Jiade Yuan; yuanjiade@fzu.edu.cn

Received 19 July 2019; Revised 23 September 2019; Accepted 30 September 2019; Published 31 October 2019

Academic Editor: Giorgio Montisci

Copyright © 2019 Jiade Yuan et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Abstract.

A compact low-profile wideband circularly polarized slot antenna for global navigation satellite system (GNSS) is presented. The antenna comprises a planar slot radiator with a coplanar waveguide (CPW) feed and a modified ring-shaped reflector to achieve unidirectional radiation. The modified reflector is an inner square patch with four slantly cut corners, a center ring, and an outer ring with a notch; they significantly reduce the separation between the antenna radiator and reflector and therefore the overall antenna height. The overall dimensions are λ₀/3 × λ₀/3 × λ₀/30 (λ₀ denotes the free space wavelength at lower frequency). The measured −10 dB bandwidth of |S₁₁|, 3-dB axial ratio (AR) bandwidth, and maximum gain are of 1.53–2.28 GHz, 1.558–1.672 GHz, and 5.87 dBi, respectively. The proposed antenna is simple without any additional feeding networks or shorting probes.

1. Introduction

Handheld terminals of global navigation satellite system (GNSS) have been popular in emergency communication, such as inshore fishery, emergency rescue, and forest fire prevention. As a key component, the antennas of the terminals not only have to be of small size, low profile, and unidirectional radiation but also must have a wide bandwidth to cover the multiple frequency bands of GNSS, such as GPS L₁ (1575.42 ± 1.023 MHz), BDS B₁ (1561.098 ± 5 MHz), and GLONASS L₁ (1602.5 ± 4 MHz) [1, 2].

Various techniques to broaden antenna bandwidth of unidirectional radiation antenna have been reported. In [3–6], parasitic patches or coupling feed structure is applied to increase bandwidth. The antenna bandwidth is increased using four parasitic strips and a capacitive-coupled feed in [3], using a Γ-shaped coupling feed structure in [4], using a tapped coupling line in [5], and using four feed strips coupling with the slot cavity in [6]. In [7], the broadband of a circularly polarized directional antenna is achieved with a circular radiating patch, a ground plane, and a novel coupling feeding network. The slot antenna’s ARBW and IBW cover the frequency band of 75.8% (1.13–2.51 GHz). In [8], a circular polarized antenna with an IBW of 0.900–2.95 GHz and ARBW of 1.0–2.87 GHz is developed. The antenna is composed of two orthogonally placed elliptical dipoles printed on both sides of a substrate. In [9], a novel wideband technique is presented based on the mode analysis and the bandwidth is expanded by loading a folded shorting pin. The operating frequency covers the band of 1.1–1.6 GHz. The antenna has an overall size of only 0.4 λ₀ × 0.4 λ₀ × 0.05 λ₀ when the relative permittivity of the substrate is 6. In [10], the antenna is proposed and has the IBW of 1.02 to 2.18 GHz and ARBW of 1.15 to 2 GHz using four claw-shaped parasitic branches. Although the abovementioned antennas have wide bandwidths and good directional radiation pattern, they have relatively larger size or complicated feeding network. As a result, they are not suitable for uses in handheld devices and for mass production.

In recent years, new antennas of simpler structures that have excellent IBW and ARBW have been reported [11–16]. In [11], a coplanar waveguide- (CPW-) fed square slot antenna is presented where an L-shaped feedline and a rectangle-shaped parasitic element is used to obtain the IBW of 63.97% and ARBW of 48.28%. In [12], by embedding a pair of inverted L-shaped strips and loading two spiral slots in the ground plane, the high IBW and ARBW of 111% and 56% are achieved, respectively. In [13], instead of using the CPW feed, the microstrip line feed is employed in the slot antenna and a transverse strip is used in the ground plane; they lead to the IBW and ARBW of 90.2% and 40%, respectively. In [14], a horseshoe-shaped slot and two L-shaped radiators are introduced, and the IBW and ARBW are of 113% (0.7–2.52 GHz) and 60.2% (3.47–6.46 GHz), respectively. Although presenting distinct advantages in IBW and ARBW, these antennas have undesirable bidirectional radiation patterns. To obtain a unidirectional radiation pattern, a simple conducting plane is placed underneath the CPW-fed slot antenna without changing its original structural dimensions [16–19]. However, they all need a relatively large distance between the radiators and the conducting planes and fail to meet the requirements of low profile.

In this paper, to overcome the shortcomings of the existing antennas for GNSS systems, we propose the integration of a broadband slot antenna with a modified reflector to form a new compact low-profile antenna that has a wide bandwidth and unidirectional radiation pattern. It has the smallest volume size among the antennas reported so far with an easy and simple coplanar waveguide (CPW) feed. An evolutional approach is presented to show the step-by-step development of the proposed antenna and its operational principles; the approach may also be used for any future developments of new antennas.

2. Configuration and Design of the Proposed Antenna

The cross-sectional view of the proposed slot antenna is shown in Figure 1(a): there are two layers of FR4 substrate separated by air; the radiator and reflector are printed on the upper and lower FR4 substrate layers, respectively.

(a)

(b)

(c)

Figure 1: The proposed antenna. (a) The cross-sectional view of the antenna, (b) the (top) view of the radiator, and (c) the (bottom) view of the reflector.

The top view of the radiator on the upper layer is shown in Figure 1(b). The antenna is fed by a 50 ohm coplanar waveguide (CPW) of a strip width of and the gaps of with the ground. The strip becomes wider as it extends further into the center of the patch with a wider width of W₁. Four grounded rectangular patches with different widths of A₁, W₃, L₂, and A₃ are etched on the top and right sides, respectively. A grounded T-shaped strip with the dimension of L₅, W₅, and W₆ is made on the left side. The geometry of the radiator (Figure 1(b) is evolved from a simple monopole antenna with a grounded loop, as explained in Section 2.1 later. The bottom view of the reflector is shown in Figure 1(c); the reflector comprises of an outer ring, a center ring, and an inner square patch with its four symmetrical corners cut slantly. A notch is etched in the outer ring.

The combination of the radiator placed on the upper layer and the reflector placed on the lower layer forms the proposed low-profile antenna with wide IBW and ARBW. The radiator and reflector should be considered and designed simultaneously because there is strong mutual coupling between them due to a small height h of about λo/30 (Figure 1(a)). In other words, the performance of the radiator is jointly affected by the radiator and the reflector. This is different from the conventional antenna presented in [16–19] where the reflector and the radiator could be designed relatively independently. In addition, through its embedded structures, such as the grounded rectangular patches and the T-shaped strip, the radiator generates orthogonal electric fields, which then lead to the circular polarization with the unidirectional radiation (achieved with the reflector).

The optimized dimensions of the proposed antenna are obtained through simulation using the ANSYS HFSS 18, and the results are listed in Table 1.

Table 1: Optimized geometric parameters (unit: mm).

Parm
Value

G₁
65.5

G₂
16

G₃
19

G₄
25.5

G₅
53

G₆
38

d₁
4.5

WS
3

LS
21.1

0.6

dcut
2.2

W₁
6.4

W₂
9

W₃
12

W₄
10

W₅
2

W₆
13

h
5

L₂
14

L₃
7

L₄
4.4

L₅
19

L₆
13.9

L₇
20

A₁
10.4

A₂
11

A₃
10

2.1. Evolution and the Design Steps of the Proposed Antenna

Six steps (Antennas I–VI) are shown in Figure 2 to explain the evolution and design process of the proposed antenna. The overall size of all the antennas is kept unchanged. The simulated results of |S₁₁|, AR at the boresight, and radiation pattern at f = 1.575 GHz of Antennas I–VI are presented in Figures 3(a)–3(c), respectively.

(a)

(b)

Figure 2: Evolution and the design steps of the proposed antenna, from Antennas I to VI. (a) The radiator on the upper layer and (b) the reflector on the lower layer.

(a)

(b)

(c)

Figure 3: Simulated reflection coefficient |S₁₁| and axial ratio of Antennas I–VI. (a) |S₁₁|, (b) axial ration, and (c) radiation pattern.

Antenna I is a conventional coplanar slot antenna, and the reflector is a square-shaped conducting plane. It has poor performances in |S₁₁|, AR, and unidirectional radiation because of the strong influence from the reflector on the radiator. Antenna II is Antenna I with the addition of a ring slot being etched in the reflector; |S₁₁| and the directivity of the radiation pattern become better. The reasons are that the etched slots reduce electromagnetic coupling between the radiator and the reflector and the resulting current distributions improve the impedance matching and unidirectional radiation. Antenna III is Antenna II with an outer ring slot and a notch being etched in the reflector that forms the inner patch, center ring, and outer ring. The notch in the reflector reduces the Q factor due to the close distance between the reflector and the SMA connector on radiator. Antenna III has better directivity than Antenna II because of the introduction of the outer slot that further optimizes the current distribution. From Antenna I to Antenna III, only the reflector is changed and the radiator is symmetric about the central strip of the CPW feed; hence, they lead to linear polarization and high AR values as shown in Figure 3(b).

To achieve the circular polarization and further improve the antenna performance, from Antenna IV to Antenna VI, the radiator is evolved, while the reflector is kept the same. Antenna IV is formed from Antenna III by embedding a T-shaped grounded strip on the left side and a rectangular grounded patch on the top left corner. As a result, the AR performance is improved significantly. The reason is that the transverse electric fields between the inverted T strip and the center strip are excited and its magnitude and phase of the electric field can generate circularly polarized radiation. Antenna V is evolved from Antenna IV by adding a rectangular patch and a stairstep-shaped grounding patches in the right side and lower right corner of the Antenna IV radiator, respectively. As shown in Figures 3(a) and 3(c), |S₁₁| and directivity improve. At last, two rectangular grounding patches are added to the lower left corner and the top of the radiator of Antenna V, respectively; they fine tune the antenna performance, and the final design of the antenna VI is formed.

2.2. Generation of Circular Polarization

The current distributions on the radiator at different time instants are simulated and studied as time t increases from to ; they are shown in Figure 4 at f = 1.575 GHz as a test point.

(a)

(b)

(c)

(d)

Figure 4: Simulated electric current distribution at 1.575 GHz. (a) t = 0, (b) t = T/4, (c) t = T/2, and (d) t = 3T/4. T is the period of the sinusoidal fields with angular frequency ω.

As seen from Figure 4(a), at t = 0, the predominant currents are J₀ on the left and right sides, and they flow in the same +y direction. As seen from Figure 4(b), at the instant of , there are four strong currents: current J₀ on the top and bottom sides that flow in the −x direction, current J₁ on the T-shaped strip that flows in the −x direction, current J₂ on the feed patch that flows in the +y direction, and current J₃ on the rectangular patch adjacent to J₂ that flows in the -y direction. J₂ and J₃ flow in the opposite directions (i.e., +y and –y directions), cancelling the far fields generated by each other; J₀ and J₁ flow in the same direction (i.e., the −x direction), reinforcing the far fields generated by each other. Therefore, at , the predominant currents are in the −x direction.

As further seen from Figures 4(c) and 4(d), the predominant currents at and are equal in magnitude and opposite in phase to the currents at and , respectively.

Consequently, the vector sum of the predominant currents flows in the anticlockwise direction with time as shown in Figure 4. Such current variations with time result in predominantly a right-hand circular polarization (RHCP) wave radiations in the z > 0 half spaces.

3. Parametric Studies

Parametric studies are performed by simulation of the antenna parameters of |S₁₁| and AR; the results allow optimization and final design of the proposed antenna. Unless indicated otherwise, only one geometrical parameter is varied each time, and the rest of the parameters are kept unchanged in the following investigations.

3.1. Dimensions of the Center Strip (or Feedline) (Ls and W₁)

|S₁₁| and AR at the boresight of the proposed antenna are affected by the dimensions of the central strip of the CPW feed. Its length and width are denoted as Ls and W₁, respectively. Simulation results are shown in Figures 5(a) and 5(b). When W₁ = 6.4 mm and Ls varies from 19.1 mm to 23.1 mm, both |S₁₁| and AR first become better and then worse. When W₁ increases from 4.4 mm to 8.4 mm with fixed Ls = 21.1 mm, |S₁₁| has slight variations while AR varies significantly and the frequency of the lowest AR value (in dB) increases. The reason is that W₁ impacts directly the distribution of transverse electric fields in the slots of the radiator and therefore affects the AR. The optimal value is Ls = 21.1 mm and W₁ = 6.4 mm with the lowest AR of 1.45 dB at the frequency of 1.575 GHz.

(a)

(b)

Figure 5: Variations of (a) |S₁₁| and (b) AR with different dimensions of center strip, Ls and W₁.

3.2. Dimensions of the T-Shaped Strip (W₆ and L₅)

|S₁₁| and AR at the boresight of the proposed antenna are affected by the lengths of the ground T-shaped strip, W₆ and L₅. Simulation results are shown in Figures 6(a) and 6(b). When L₅ = 19 mm and W₆ increases from 12 mm to 14 mm, |S₁₁| has slight variation while AR shows visible changes. This indicates that AR is more sensitive to W₆; the reason is that the transverse electric field between the T-shaped strip and the center feed strip is sensitive to the gap width (determined by W₆) as it forms the one of the axial polarizations for the circular polarization. For the similar reason, when L₅ increases from 16 mm to 22 mm with the fixed W₆ = 13 mm, |S₁₁| varies slightly and AR shows visible variations.

(a)

(b)

Figure 6: Variations of (a) |S₁₁| and (b) AR with different dimensions of the T-shaped strip, W₆ and L₅.

3.3. Dimension of the Inner Square Patches of the Reflector (G₂)

Figures 7(a) and 7(b) show the simulation results of |S₁₁| and AR and realized gain of RHCP with the variation of G₂, respectively. It can be observed from Figure 7(a) that the resonant frequency is shifted to the lower value with the increasing of G₂ from 15 mm to 17 mm. The reason can be explained that the equivalent capacitor between the radiator and reflector is increased because the increasing of G₂ means the decreasing of the width of the inner slot of the reflector. Meanwhile, the AR changes to better then worse (see AR curves in Figure 7(b)). The realized gains of RHCP also vary with the increasing of G₂ and appear lower values when G₂ = 17 mm, as shown the realized gain curves in Figure 7(b). The main reason is due to the poor AR performance when G₂ = 17 mm.

(a)

(b)

Figure 7: Variations of (a) |S₁₁| and (b) AR and realized gain with different dimensions of the inner square patches, G₂.

3.4. Dimensions of the Center Ring of the Reflector (G₃ and d₁)

G₃ and d₁ (shown in Figure 1(b)) represent the relative position and width of the center ring in the reflector, respectively. Figures 8(a) and 8(b) show |S₁₁|and AR in the operating frequency rang with different G₃ and d₁. It can be observed that |S₁₁| has slight variations while AR has visible changes with the increase of G₃ from 18 mm to 20 mm and the increase of d₁ from 3.5 mm to 5.5 mm. This indicates that AR is more sensitive to the variation of the relative position and width of the center ring.

(a)

(b)

Figure 8: Variations of (a) |S₁₁| and (b) AR with different dimensions of the center ring of the reflector, G₃ and d₁.

3.5. Dimensions of the Outer Ring of the Reflector (G₁ and G₄)

|S₁₁| and AR at the boresight of the proposed antenna with varying lengths of the outer ring of the reflector, G₁ and G₄, are computed, and the results are shown in Figures 9(a) and 9(b). When G₁ increases from 60.5 to 70.5 mm and G₄ increases from 24.5 mm to 26.5 mm, |S₁₁| has slight variations while AR shows visible variations. The optimal values for small AR and antenna size are G₁ = 65.5 mm and G₄ = 25.5 mm.

(a)

(b)

Figure 9: Variations of (a) |S₁₁| and (b) AR with different dimensions of the outer ring of the reflector, G₁ and G₄.

Based on the above studies, the final geometrical parameters of the proposed antenna are listed in Table 1.

4. Experimental Results and Discussion

Based on the optimized parameters listed in Table 1, a prototype of the proposed antenna is fabricated, as shown in Figures 10(a)–10(c) for top, bottom, and side views, respectively, where the two layers are separated and supported with four plastic posts.

(a)

(b)

(c)

Figure 10: The fabricated prototype of the proposed antenna. (a) Top view, (b) bottom view, and (c) side view.

The prototype antenna was measured with Agilent E5071Cvector network analyzer for |S₁₁| and for radiation pattern and AR in our anechoic chamber. The results are shown in Figure 11 It can be seen that the measured results agree reasonably well with the simulated ones. The measured |S₁₁| frequency band of −10 dB is 1530 MHz to 2280 MHz which covers the full operating bandwidth of BD2 B₁, GPS L₁, and GLONASS L₁.

Figure 11: Simulated and measured |S₁₁| of the proposed antenna.

The simulated and measured AR and gain are shown in Figure 12(a). The measured 3 dB AR frequency ranges from 1558 MHz to 1672 MHz in RHCP. The measured results agree reasonably well with the simulated results. The measured gain varies with frequency and the gain is 5.26, 5.78, and 5.12 dBi at the center frequency of BD2 B₁ (1561 MHz), GPS L₁ (1575 MHz), and GLONASS L₁ (1602 MHz), respectively. The maximum measured gain is 5.87 dBi in the whole operating frequency band. The simulated and measured radiation efficiencies are shown in Figure 12(b). As can be seen, the measured radiation efficiencies are around 80%, while the simulated ones are around 90%. The discrepancy between simulations and measurements are mainly due to the errors of the conductivity of conductors and dielectric loss angle of tangent of the antenna between the theoretical values and the true values.

(a)

(b)

Figure 12: Measured and simulated results of (a) gain and AR at the boresight and (b) radiation efficiency.

The normalized RHCP and LHCP radiation patterns of the proposed antenna at the frequency of 1.561, 1.575, and 1.602 GHz in the xz plane and yz plane are simulated and measured, respectively, and the results are shown in Figure 13. In all cases, the measured and simulated results have good corroborations.

(a)

(b)

(c)

(d)

(e)

(f)

Figure 13: Simulated and measured radiation patterns in the xz plane and the yz plane at (a) f = 1.561 GHz, (b) f = 1.575 GHz, and (c) f = 1.602 GHz.

Table 2 shows the comparison of the antenna size and performance between the proposed antenna and other recently published works. Here, we introduce the ratios of antenna volume size to relative bandwidth of |S₁₁| and AR, RVB1, and RVB2, respectively; they are to measure the relative sizes of the antennas. Such ratio definitions for comparisons are justifiable and representative since the antenna size is proportional to the frequency it operates at. The smaller the ratio is, the better performance the antenna has in terms of the size.

Table 2: Comparison of the antenna size and performance between this work and other recently published works.

Ref. no
Height (λ₀)
Area (λ₀²)
IBW (%)
RVB1 (×10⁻²)
ARBW (%)
RVB2 (×10⁻²)

[4]
0.13
0.45 × 0.45
43.8
6.01
42.6
6.18

[5]
0.042
0.88 × 0.88
8.52
38.17
7
46.4

[6]
0.096
0.6 × 0.6
40.0
8.64
34.0
10.15

[7]
0.09
0.7 × 0.8
97.8
5.15
83.8
6.02

[15]
0.076
0.44 × 0.44
33.5
4.39
27.2
5.41

[16]
0.22
0.54 × 0.54
52.6
12.2
34.4
18.6

[17]
0.22
0.85 × 0.85
109.6
14.6
60.9
26.3

This work
0.033
0.3 × 0.3
41.8
0.83
7.1
4.9

Height = total profile height; area = area of the reflector; IBW = −10 dB |S₁₁| relative BW; RVBR1 = ratio of antenna volume size to relative bandwidth of the |S₁₁|; ARBW = 3 dB AR relative BW; RVB2 = ratio of antenna volume size to relative bandwidth of the AR.

As can be seen from Table 2, the proposed antenna has the smallest height of 0.033λ₀ for profile and transverse area of 0.33 × 0.33 λ₀², with the smaller ratio of 0.83 for RVB1 and 4.9 for RVB2. Therefore, it is the compact antenna with better performance of IBW and ARBW.

5. Conclusions

The CPW feed slot antenna is combined with the modified reflector to develop a new compact low-profile antenna with unidirectional radiation for GNSS application. The modified reflector contains an inner square patch with four slantly cut corners, a center ring, and an outer ring with a notch and has a small transverse area of 0.33 × 0.33 λ₀². The proposed design can significantly reduce the separation gap between the radiator and the reflector and then result in a smaller antenna profile of about λ₀/30. It has a wide IBW and ARBW of 41.8% and 7.1%, respectively, and meanwhile, the peak measured gain can reach 5.87 dBi in the whole operating frequency band. In addition, its structure is easily fabricated since it has a simple CPW feed and does not need additional feeding networks and shorting probes. It may also be used conveniently as an antenna array element because of its side-fed structure, low profile, and small size.

Data Availability

The data used to support the findings of this study are included within the article.

Conflicts of Interest

The authors declare that they have no conflicts of interest.

Acknowledgments

This work was supported by the Natural Science Foundation of Fujian Province of China under Grant no. 2019J01638 and the Science and Technology Project Plan of Fuzhou of China under Grant no. 2018-G-89.

References

J. J. H. Wang, “Antennas for global navigation satellite system (GNSS),” Proceedings of the IEEE, vol. 100, no. 7, pp. 2349–2355, 2012.

J. Yuan, J. Zheng, and Z. D. Chen, “A compact meandered ring antenna loaded with parasitic patches and a slotted ground for global navigation satellite systems,” IEEE Transactions on Antennas and Propagation, vol. 66, no. 12, pp. 6835–6843, 2018.

J. Wu, Y. Yin, Z. Wang, and R. Lian, “Broadband circularly polarized patch antenna with parasitic strips,” IEEE Antennas and Wireless Propagation Letters, vol. 14, pp. 559–562, 2015.

H. Zhang, Y. Guo, and G. Wang, “A wideband circularly polarized crossed-slot antenna with stable phase center,” IEEE Antennas and Wireless Propagation Letters, vol. 18, no. 5, pp. 941–945, 2019.

Q.-S. Wu, X. Zhang, and L. Zhu, “A feeding technique for wideband CP patch antenna based on 90° phase difference between tapped line and parallel coupled line,” IEEE Antennas and Wireless Propagation Letters, vol. 18, no. 7, pp. 1468–1471, 2019.

K. Luo, B. Chen, and W.-P. Ding, “Meander line coupled cavity-backed slot antenna for broadband circular polarization,” IEEE Antennas and Wireless Propagation Letters, vol. 14, pp. 1215–1218, 2015.

Q. Liu, Y. Li, Z. Mo, and Y. Liu, “Compact broadband circularly-polarized directional universal GNSS antenna with symmetric radiation pattern and stable near-zenith coverage,” IET Microwaves, Antennas & Propagation, vol. 11, no. 5, pp. 657–663, 2017.

L. Zhang, S. Gao, Q. Luo et al., “Single-feed ultra-wideband circularly polarized antenna with enhanced front-to-back ratio,” IEEE Transactions on Antennas and Propagation, vol. 64, no. 1, pp. 355–360, 2016.

C. Sun, Z. Wu, and B. Bai, “A novel compact wideband patch antenna for GNSS application,” IEEE Transactions on Antennas and Propagation, vol. 65, no. 12, pp. 7334–7339, 2017.

L. Wang, Z. Weng, Y.-C. Jiao, W. Zhang, and C. Zhang, “A low-profile broadband circularly polarized microstrip antenna with wide beamwidth,” IEEE Antennas and Wireless Propagation Letters, vol. 17, no. 7, pp. 1213–1217, 2018.

R. K. Saini and S. Dwari, “CPW-fed broadband circularly polarized rectangular slot antenna with l-shaped feed line and parasitic elements,” Microwave and Optical Technology Letters, vol. 57, no. 8, pp. 1788–1794, 2015.

P. Sadeghi, C. Ghobadi, and J. Nourinia, “Square slot antenna with two spiral slots loaded for broadband circular polarization,” Electronics Letters, vol. 52, no. 10, pp. 787-788, 2016.

M. S. Ellis, Z. Zhao, J. Wu, X. Ding, Z. Nie, and Q.-H. Liu, “A novel simple and compact microstrip-fed circularly polarized wide slot antenna with wide axial ratio bandwidth for C-band Applications,” IEEE Transactions on Antennas and Propagation, vol. 64, no. 4, pp. 1552–1555, 2016.

R. Xu, J.-Y. Li, J. Liu, S.-G. Zhou, Z.-J. Xing, and K. Wei, “A design of dual-wideband planar printed antenna for circular polarization diversity by combining slot and monopole modes,” IEEE Transactions on Antennas and Propagation, vol. 66, no. 8, pp. 4326–4331, 2018.

M. F. Farooqui and A. Kishk, “3-D-Printed tunable circularly polarized microstrip patch antenna,” IEEE Antennas and Wireless Propagation Letters, vol. 18, no. 7, pp. 1429–1432, 2019.

S. Pan, S. Jia-Yi, and T. Po-Jen, “Circularly polarized square slot antenna with a largely enhanced axial-ratio bandwidth,” IEEE Antennas and Wireless Propagation Letters, vol. 11, pp. 969–972, 2012.

G. Li, H. Zhai, L. Li, and L. Changhong, “A nesting-L slot antenna with enhanced circularly polarized bandwidth and radiation,” IEEE Antennas and Wireless Propagation Letters, vol. 13, pp. 225–228, 2014.

S. H. Yeung, K. F. Man, and W. S. Chan, “A bandwidth improved circular polarized slot antenna using a slot composed of multiple circular sectors,” IEEE Transactions on Antennas and Propagation, vol. 59, no. 8, pp. 3065–3070, 2011.

R. Xu, J.-Y. Li, K. Wei, and G.-W. Yang, “A broadband slot antenna with unidirectional circularly polarized radiation patterns,” IEEE Antennas and Wireless Propagation Letters, vol. 16, pp. 317–320, 2017.