A Dual-Band SiGe HBT Frequency-Tunable and Phase-Shifting Differential Amplifier Employing Varactor-Loaded, Stacked LC Resonators

A dual-band SiGe HBT frequency-tunable and phase-shifting differential amplifier has been developed for the future active phased array antennas with a multiband, multibeam, and multitarget tracking operation. The amplifier uses varactor-loaded, stacked LC resonators in the design of the output circuit in order to provide frequency-tunable and phase-shifting capabilities for dual frequencies. By utilizing the varactor-loaded LC resonator, which has a variable resonant frequency and a large insertion phase variation, frequency-tunable and phase-shifting performances become available. Moreover, by using the stacked configuration, the frequency and insertion phase can be varied independently for dual frequencies. A dual-band SiGe HBT differential amplifier has achieved a lower-frequency tuning range of 0.56 to 0.7 GHz for a higher fixed frequency of 0.97 GHz as well as a higherfrequency tuning range of 0.92 to 1.01 GHz for a lower fixed frequency of 0.63 GHz. A lower-frequency phase variation of 99◦ and a higher-frequency phase variation of 90.3◦ have been accomplished at 0.63 and 0.97 GHz, respectively. This is the first report on the dual-band differential amplifier with frequency-tunable and phase-shifting capabilities.


Introduction
Recent and future wireless systems require a wide range of data rates over several frequency bands, and thus adaptive and reconfigurable transceivers become necessary to support high performance and flexibility [1]. Meanwhile, microwave and millimeter-wave sensors and radar systems require multiband, multibeam scanning phased array antennas as well as a multi-target tracking capability [2]. To address these requirements, multifunction capabilities are strongly required. Especially multiband amplification, frequency tunability, gain and phase control are crucial for realizing adaptive and reconfigurable phased array systems. Most of the traditional multiband amplifiers, however, provide a single function [3]. Meanwhile, the authors have presented multiband amplifiers with frequency-tunable as well as gain control capabilities for bandpass and bandstop types [4,5]. To enhance the operational capability, a novel design approach is proposed in this paper for the multiband low-noise differential amplifier with frequencytunable and phase-shifting capabilities. In order to realize both frequency-tunable and phase-shifting performances, varactor-loaded LC resonators are used in the design of the output circuit of the differential amplifier. LC resonators are widely used in the reflection type phase shifter because they provide a full 360 • phase shift [6][7][8]. With the use of the parallel LC resonator in the output shunt circuit, frequency-tunable and phase-shifting performances with high gain can be achieved at around resonant frequencies.
In addition, by cascading the parallel LC resonators in a stacked form, frequency and insertion phases can be varied independently for multiple frequencies. As a multiband phase shifter, the resonator-switching or diplexer method has been proposed for the loaded-line type [9], reflection type [10], and LPF/HPF type [11]. These phase shifters, however, produce additional losses due to the switch or diplexer.
Since the stacked LC resonator does not require additional switches or diplexers, a low insertion loss and a large phase variation can be expected. To make sure the usefulness of the novel design approach proposed in this paper, the dualband SiGe HBT differential amplifier with frequency-tunable and phase-shifting capabilities has been actually designed, fabricated, and tested.

Circuit Design
A schematic diagram of the dual-band frequency-tunable and phase-shifting differential amplifier is shown in Figure 1. It employs varactor-loaded, stacked LC resonators in the design of the output circuit to achieve frequency-tunable and phase-shifting capabilities for dual frequencies. L 1 , L 2 , C 1 , and C 2 are inductors and capacitors consisting of LC resonators. C V 1 and C V 2 are a variable capacitance of the varactor diode which is controlled by V C1 and V C2 . C C is a decoupling capacitor having a large capacitance value. Resonant frequencies f 1 and f 2 are given as follows: When two parallel LC resonators are cascaded in a stacked form, the following bandstop frequency f 3 is generated. Consider Now f 1 < f 2 (L 1 C 1 > L 2 C 2 ) then f 3 is necessarily sandwiched between f 1 and f 2 as (4) C V 1 and C V 2 are carefully designed in accordance with the target frequency range and phase-shifting amount. In order to realize C V 1 and C V 2 , several varactor diodes with a fixed capacitance ratio are connected in series or parallel.
A locus of the reflection coefficient Γ L of the varactorloaded, stacked LC resonator is graphically shown in Figure 2. As the frequency increases from zero to f 1 , Γ L moves from the short to the open-circuited point P. Since the LC resonator is actually lossy, the point P has finite impedance. Then as the frequency increases up to f 3 , Γ L moves to the series resonant point q where the impedance is not ideally short due to the lossy circuit. As the frequency increases from f 3 to f 2 , Γ L comes back to the point P again. Finally Γ L goes to short. At around the resonant frequencies f 1 and f 2 , the insertion phase drastically varies from +180 • to −180 • . Therefore, with the precise control of V C1 and V C2 , the insertion phase can be tuned. In addition, at the resonant frequencies f 1 and f 2 , a peak gain of the amplifier can be obtained. The mechanism is depicted graphically in Figure 3 by using a frequency response of the amplifier.
It is clearly demonstrated that the frequency can be varied with the coarse control of V C1 and V C2 and that the insertion stacked LC-resonator Varactor-loaded, In− In+   phase can be also varied with the fine control of V C1 and V C2 . f 1 , f 2 , and f 3 are calculated by using from (1) to (3) for the circuit element values listed in Table 1. The circuit element values were chosen to meet the condition of (4). When C V 1 varies from 6 to 15 pF, C V 2 has a fixed value of 6 pF. Meanwhile, when C V 2 varies from 6 to 15 pF, C V 1 has a fixed value of 15 pF. The calculated results are shown in Figures 4 and 5, respectively. It is noted that f 1 and f 2 can be tuned independently and that f 3 is necessarily sandwiched between f 1 and f 2 .

Circuit Fabrication
A photograph of the dual-band frequency-tunable and phase-shifting differential amplifier is shown in Figure 6. The differential amplifier was fabricated on the FR-4 substrate with a dielectric constant of 4.5. 0.35 μm SiGe HBT with an f t of 25 GHz (Toshiba MT4S102T), Si varactor diode with a capacitance ratio of 2.5 : 1 (Toshiba 1SV279), 1005-type chip resistors, and inductors and capacitors are mounted on the substrate by soldering. The circuit size is 16 × 16 × 1.2 mm 3 .

Circuit Performance
4.1. Frequency-Tunable Performances. Measured gain, input and output return losses for a fixed V C1 (C V 1 ) and a variable V C2 (C V 2 ) are shown in Figures 7, 8, and 9, respectively. As a coarse tuning, V C2 (C V 2 ) was varied from 6 to 18 V with 1 V step since the anode voltage is equal to V CC of 6 V. The maximal reverse voltage is 15 V. f 2 moves from 0.73 to 1.31 GHz for a fixed f 1 of 0.56 GHz. The gain varied from 0 to 7.2 dB for f 2 and keeps a constant value of 8.9 dB for f 1 . V CC and V C1 were 6 V. Since the LC resonators are used in the output circuit, the output return loss is varied with the  resonant frequency, but the input return loss keeps constant. Since the amplifier employed a reflective match approach for low-noise figure, the input and output return losses were less than 10 dB, which can be greatly improved by employing the lossy match approach using resistors [4]. In [4], the input and output return losses can be improved up to greater than 10 dB across the same frequency band by using the lossy match approach. This approach, however, provides a low gain and a poor noise figure. Thus, the reflective match approach was chosen. Meanwhile, the measured gain, input and output return losses for a variable V C1 (C V 1 ) and a fixed V C2 (C V 2 ) are shown in Figures 10, 11, and 12, respectively. In a similar way, V C1 (C V 1 ) was varied from 6 to 18 V with 1 V step. f 1 moves from 0.57 to 0.97 GHz and f 2 from 1.31 to 1.34 GHz. The gain varied from 8.7 to 13.1 dB for f 1 and 1.3 to 6.5 dB for f 2 . V CC and V C2 were 6 V. If the varactor-loaded LC resonator shows an ideally open-circuited performance, f 1 and f 2 can be controlled independently. But actually the gain and frequency shift at f 2 was affected to a certain extent by controlling V C1 (C V 1 ). This is because the parasitic resistance and low-Q factor of the varactor diode (C V 1 ) have made a serious effect on the gain and frequency at f 2 . Therefore, V C2 (C V 2 ) in addition to V C1 (C V 1 ) must be adjusted simultaneously for keeping f 2 constant. Moreover,  as f 1 comes close to f 2 , the bandwidth at f 1 becomes wider, and the gain decreases in Figure 10. This is considered to be the same reason of the parasitic resistance and low-Q factor of the varactor diode. In Figures 7 and 10, the 3rd gain peaking can be seen around 1.6 GHz. This is mainly due to the fact that the LC resonator in Figure 1 shows a dual-band resonation due to the lead inductance of the varactor diode.

Phase-Shifting Performances.
Measured phase variations for a fixed V C1 (C V 1 ) and a variable V C2 (C V 2 ) are shown in Figure 13. As a fine tuning, V C2 (C V 2 ) was varied from 10  to 12 V with 0.2 V step. f 2 moves from 0.92 to 1.01 GHz for a fixed f 1 of 0.63 GHz. The phase variations at 0.97 GHz and 0.63 GHz were 90.3 • and 2.5 • , respectively. V CC and V C1 were 6 V and 7 V, respectively. The gain variation, however, was 7 dB at 0.97 GHz. The gain variation can be improved by using a lossy match approach, which in turn degrades a phase-shifting amount; that is, there exists a design tradeoff between phase and gain variations. As one approach to address this problem, an additional variable gain amplifier (VGA) with a constant phase is utilized to compensate for the gain variation. Meanwhile, the measured phase variations for a fixed V C2 (C V 2 ) and a variable V C1 (C V 1 ) are shown in Figure 14. In a similar way, V C1 (C V 1 ) was varied from 6 to 8 V with 0.2 V step.     Figure 15 for a fixed V C1 (C V 1 ) and a variable V C2 (C V 2 ) and in Figure 16 for a variable V C1 (C V 1 ) and a fixed V C2 (C V 2 ). V CC , V C1 , and V C2 were 6 V, 7 V and 9-13 V in Figure 15. Meanwhile, V CC , V C1 , and V C2 were 6 V, 5-9 V, and 11 V in Figure 16. The measured noise figures were less than 4.1 dB at 0.63 GHz and less than 5.3 dB at 0.97 GHz in Figure 15. Meanwhile, the measured noise figures were  3 . IIP 3 performances were measured with two tones of a 10 MHz separation for a fixed V C1 (6 V) and a variable V C2 (6-18 V) as well as a variable V C1 (6-18 V) and a fixed V C2 (6 V). V CC was 6 V. The amplifier has shown the minimum IIP 3 of −8.7 dBm for a fixed V C1 (6 V) and a variable V C2 (6-18 V) as well as −6.5 dBm for a variable V C1 (6-18 V) and a fixed V C2 (6 V). A total collector current I C appeared in all measurements and ranged from 9 to 12 mA.

Conclusions
The dual-band SiGe HBT frequency-tunable and phaseshifting differential amplifier has been presented for use in the future active phased array antennas with a multiband, multibeam, and multi-target tracking operation. To realize multifunction capabilities including frequency-tunable and phase-shifting capabilities, the novel design approach to incorporate varactor-loaded, stacked LC resonators with different resonant frequencies into the circuit design has been proposed. Then to make sure the usefulness of the design approach, the dual-band frequency-tunable and phase-shifting differential amplifier using SiGe HBT and Si  varactor diode have been actually designed, fabricated, and tested. It is confirmed from the measured results that the design approach proposed in this paper would be one of the candidates for the multifunctional amplifier for use in the current and future wireless, aerospace, microwave, and millimeter-wave sensors and radar systems.