International Journal of Antennas and Propagation

Volume 2013 (2013), Article ID 179296, 6 pages

http://dx.doi.org/10.1155/2013/179296

Research Article

Design of a Printed Dipole Antenna Array for a Passive Radar System

Peter Knott

Department of Antenna Technology and Electromagnetic Modelling, Fraunhofer Institute for High Frequency Physics and Radar Techniques FHR, 53343 Wachtberg, Germany

Received 20 February 2013; Accepted 19 April 2013

Academic Editor: Càndid Reig

Copyright © 2013 Peter Knott. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Abstract.

Passive radar (or Passive Coherent Localisation) is an advancing technology for covert operation. The signal transmitted from sources of opportunity such as radio or TV stations is used as illumination for a certain area of interest. Part of the transmitted signal is reflected by radar targets, for example, moving objects such as vehicles or aircraft. Typical radar parameters are derived from the comparison between the direct line-of-sight from the transmitter and the signal scattered from the target object. Such systems are an attractive addition to existing active radar stations because they have the potential to discover low-flying and low-observable targets and no active radar transmitter is required. Printed dipole antennas are very attractive antenna elements for such systems because of their easy fabrication, low-cost, polarisation purity, and low-profile properties. The present paper describes the design of an antenna array using printed dipole elements with flared arms for a passive radar system operating in the GSM900 frequency range. Isolated antenna elements and a small uniform linear antenna array were designed and optimised using computational electromagnetic methods. Several prototypes have been fabricated on conventional microwave PCB substrate material. Preliminary measurement results for antenna matching and far-field radiation patterns are shown.

1. Introduction

Passive radar, also referred to as Passive Coherent Localisation (PCL), making use of RF sources of opportunity such as radio or TV broadcasting stations and cellular phone network base Stations is an advancing technology for localisation of radar targets and covert operation. Such systems are an attractive addition to existing active radar stations because they have the potential to discover low-flying and low-observable targets and no active radar transmitter is required. Different passive radar systems are currently being developed, such as the CORA (Covert Radar) experimental system featuring a multichannel digital radar receiver and a circular antenna array [1, 2].

Instead of actively transmitting radar pulses, Passive Radar, also known as Passive Coherent Localisation, uses signals from available transmitters (sources of opportunity) for radar operation. Echoes from moving objects can thus be detected as targets. The principle of bistatic radar operation is illustrated in Figure 1: the signals transmitted from sources-of-opportunity such as radio, TV or mobile communication base stations are used as illumination for a certain area of interest. Part of the transmitted signal is reflected by radar targets, for example, moving objects such as vehicles or aircraft. From the comparison between the direct line-of-sight (LoS) signal from the transmitter and the signal scattered from the target object, typical radar parameters such as direction, range, or velocity may be derived at the receiver, which is typically built as a multichannel system. The subject of Passive Bistatic Radar (PBR) has been comprehensively covered in different textbooks and journals, for example, [3–6].

Figure 1: Signals transmitted from sources of opportunity are used as illumination to detect radar targets. Typical radar parameters such as direction, range, or velocity may be derived from the comparison between the direct line-of-sight (LoS) signal from the transmitter and the signal scattered from the target object.

The widespread existence of digital wireless communication installations and digital broadcasting stations makes the application of passive radar using these stations as illuminators attractive. Passive radar systems using analogue FM radio transmitters have been designed and systems are now commercially available. Currently there is a widespread interest in developing DAB/DVB passive radar systems because digitally coded signals have an autocorrelation function practically independent of the information content. Digital waveforms have a time-invariant bandwidth, which is typically wider than FM radio waveforms. They are, therefore, capable of yielding finer range resolution. An experimental system for verifying the feasibility of passive radar in the DVB-T frequency range has been developed at Fraunhofer FHR and is currently operated for measurements and optimisation of signal processing. Experimental systems for verifying the feasibility of GSM passive radar are currently being developed and built at the Fraunhofer Institute for Communication, Information Processing and Ergonomics FKIE [7, 8]. The experimental systems comprise uniform linear arrays (ULA) with 32 elements and with 16 elements resulting in an azimuth beamwidth of 3.2° and 6.4°, respectively. The larger array allows for the application of powerful subspace projection methods to cancel the direct LoS path signal. The smaller antenna has been reduced in size towards a more operational system and operates on a 16-element array.

2. Antenna Design

Printed dipole antennas are very attractive antenna elements in the RF and microwave frequency region because of their easy fabrication, low cost, polarisation purity, and low-profile properties. Theory and design principles of straight and flared printed dipole antennas and antenna arrays for single and dual polarization along with feeding and broadband matching techniques have been treated in several publications [9–11]. They can be designed for a relative bandwidth of up to approximately 10% without applying any additional loading or impedance transformation techniques and many examples of industrial application can be found.

Antenna arrays for multichannel passive radar systems can be arranged in different configurations, depending on the geometrical alignment of transmitter location, receiver location, and the area under surveillance. Several array geometries have already been explored and used in demonstrator systems, including circular antenna arrays. For the typical scenario of the present application, a uniform linear array (ULA) seemed to be the most appropriate and practical choice of geometry. The antenna elements used in such a linear array may be fabricated and aligned in many different ways according to the system specification. Two examples are shown in Figure 2: several antenna elements of a linear array may be printed on a single printed circuit board (PCB) substrate in a horizontal alignment (Figure 2(a)). In this case, the mutual electromagnetic coupling is not so strong because the antenna elements are located in the radiation minima of their neighbours [12]. On the other hand, the spacing between neighbouring elements is dependent on the dielectric permittivity of the substrate and may well exceed half a wavelength. If the element spacing becomes too large, array grating lobes may occur for large scan angles. Alternatively, each element may be fabricated on an individual piece of PCB and aligned in a vertical fashion such that all dipole axes are in parallel (see Figure 2(b)). Thus, also relatively small interelement distances may be realised while the influence of mutual coupling increases at the same time.

Figure 2: Alignment of antenna elements in linear arrays (a) horizontal on single PCB substrate layer and (b) vertical on individual PCBs for each antenna elements.

For an existing passive radar demonstrator systems with receivers operating in the GSM 1800 frequency band, new components, and additional antennas for the GSM 900 range operating at centre frequency  MHz are currently being developed. For the system currently under development, the antenna elements have to fulfil the following design criteria: they should be horizontally polarised and operate in the required frequency range of f = 900–1,000 MHz. Their dimensions should be as small as possible and allow for an interelement spacing smaller than half a wavelength in free space. Therefore, a design with flared dipole arms as shown in Figure 3 was chosen. The dipole arms with a flare angle of 45° are printed on opposite sides of the PCB and connected to a feeding balanced double transmission line. The length and characteristic wave impedance of this feeding section is used as a quarter-wavelength transformer and optimised to directly match the 50 Ω microstrip transmission line at the antenna port where a coaxial connector is attached. The electrical equivalent circuit (EEC) of a single antenna element and its matching section are shown in Figure 5. Due to the relatively small fractional bandwidth of the system, it was possible to omit any additional BALUN or impedance transformer sections in order to keep the PCB size as small as possible. This results in a narrow-band matching of the antenna input impedance to the characteristic impedance of the system which is perfect only at the designed frequency of resonance. It will be shown later, however, that the resulting mismatch is tolerable over the frequency range of operation. As a consequence of the nonsymmetric design, an increased level of cross-polarisation was accepted.

Figure 3: Antenna element prototype with flared dipole arms printed on PCB with coaxial connector attached.

3. Electromagnetic Simulation and Measurements

The antenna dimensions and design parameters were analysed using a full-wave electromagnetic analysis software [13], both for an isolated antenna element and for an antenna array of finite size with the prescribed element spacing of 160 mm (equivalent to 0.53 λ at the upper frequency limit). In order to include the interelement mutual coupling effects, a realistic scenario with a small linear array was used consisting of five equally spaced antenna elements. To increase the forward directivity, avoid backward radiation, and protect the electronic components from electromagnetic interference, the antennas are mounted and operate in front of a conducting backplane which is also taken into account in the simulations as a perfectly electrically conducting (PEC) plane. The radiation pattern of a single dipole antenna element is mirrored at this conducting plane such that the back-lobe is removed and the forward directed lobe has a higher gain (ideally +3 dB). However, since the ground plane is not perfectly conducting and has a finite size only, the practical improvement in gain is smaller.

For experimental verification, several antenna prototypes have been built up on commercial dielectric substrate material [14] with a relative dielectric constant of 2.2 and a thickness of 0.81 mm. One sample of the flared printed dipole antenna element is shown in Figure 3. The overall size of the printed circuit board is 137.5 mm (length) and 108.5 mm (width). Each antenna element is fed via a transition from a coaxial connector to a 50 Ω microstrip transmission line mounted on a short piece of rectangular ground plane with limited size. Among the design goals of the system was a small interelement distance (to avoid ambiguities/grating lobes), low mutual coupling between neighbouring elements, and small increase of cross-polarization isolation which are somewhat conflicting properties. Therefore, a parametric study of the influence of the flare angle on the input reflection coefficient and mutual coupling has been conducted, as shown in Figure 4. The interelement distance of 160 mm and the dimensions of the dipole arms (width and length) were fixed during these simulations. For a flare angle of 45°, the mutual coupling drops below −15 dB over the full frequency range while the width of the PCB can be reduced to 108 mm. It has to be noted, however, that due to changes in the parasitic capacity between the dipole arms and the feeding transmission line, the antenna input reflection coefficient increases and the frequency of resonance is shifted for different values of α. Thus, for flare angles other than 45°, a small readjustment of the dipole antenna geometry becomes necessary. The different geometric antenna parameters (e.g., dipole arm width, bending angle, length, and width of the impedance transformer section, etc.) have been carefully optimized with respect to the specifications and constraints described above. The dimensions and antenna parameters found during the optimization are listed in Table 1.

Table 1: Dipole antenna element design parameters and dimensions.

Parameter
Value

PCB length
137.5 mm

PCB width
108.5 mm

Dipole arm length
143 mm

Dipole arm flare angle
45°

Design frequency of resonance
940 MHz

Interelement spacing
160 mm

(a)

(b)

Figure 4: Simulation results for a parametric study of the influence of the flare angle α (off the normal dipole axis) for a given interelement distance of 160 mm and fixed dimensions of the dipole arms (a) magnitude of the input reflection coefficient and resonance frequency (b) magnitude of the mutual coupling between two neighbouring antenna elements.

Figure 5: Electrical equivalent circuit—the input impedance of the antenna is matched to the system impedance using matching microstrip line section as a quarter wavelength transformer.

After fabrication, the antenna elements have been assembled in a linear array of the same size and spacing according to the simulations. The S-parameters of the centre element and its neighbour have been measured using a network analyser. A comparison between simulation and experimental results for input reflection coefficient () is shown in Figure 6. A reflection coefficient of −10 dB or below is achieved for the frequency range from 880 MHz to 1020 MHz corresponding to a relative bandwidth of 14.7%. A small displacement of the measurement curve against the simulation results is observed and may be attributed to fabrication tolerances, the finiteness of the metallic backplane in the experiment, and the influence of the coaxial connector which are not included in the electromagnetic simulation. Figure 7 shows the three-dimensional radiation pattern of the centre antenna element inside the array for the frequency of  MHz as an example. In this illustration, the element is radiating mainly into the z-direction while the dipole axis is oriented along the -axis. A 2-dimensional cross-cut of the directivity simulation results in the xz-plane (°) is given in Figure 8. Due to the asymmetry of the design, the radiation pattern is not symmetric to the yz-plane and the direction of maximum directivity (6.26 dBi) is slightly shifted away from the zenith (z-axis).

Figure 6: Antenna element matching—comparison of measured and simulated (CST Microwave Studio) input reflection coefficient at the antenna element embedded inside the 5-element antenna array.

Figure 7: Simulated results for 3D antenna radiation pattern (directivity) of a single antenna element embedded inside the 5-element array at  MHz.

Figure 8: Simulated antenna radiation pattern (directivity) in -plane () at  MHz.

A larger antenna array with multiple horizontally polarized printed dipole antenna elements is currently being fabricated. A special mounting facility was designed such that the antenna elements can be rotated around their longitudinal axis before they are fixed inside the metallic backplane. This flexible fixture gives an additional degree of freedom to the array design and opens up the possibility to compensate for mutual coupling effects and/or cross-polarisation by rotating the elements to an optimum angle. Figure 9 shows a picture of a previously built antenna array for a different frequency range using a flared printed dipole element type and an identical fixture for illustration. Before the system can go into operation, the array manifold including the embedded radiation patterns of all antenna elements has to be measured and the individual channels of the digital multichannel receiver have to be calibrated.

Figure 9: Picture of the receiving antenna array with rotatable antenna mounting.

4. Conclusions

The design of an antenna array using printed dipole elements with flared arms for a passive radar system operating in the GSM 900 frequency range has been presented. Printed dipole antennas have been selected as antenna elements because they well suited for the given specifications and easy to design and fabricate. Single antenna elements and a small uniform linear antenna array were designed and optimised using computational electromagnetic methods. It has been found that on the dielectric substrate material used and the desired frequency range of operation, an element size of 137.5 mm by 108.5 mm (PCB size), a dipole arm length of 143 mm, and a flare angle of 45° delivered the optimum performance in terms of antenna matching, mutual coupling, and cross-polarization isolation. Several prototypes have been fabricated on conventional microwave PCB substrate material. Preliminary measurement results for antenna matching and far-field radiation patterns are in good agreement with electromagnetic simulations. Future work will include the fabrication and measurement of a full-scale antenna array. The antenna will have to be integrated into the multichannel receiver system and carefully calibrated before operation.

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