International Journal of Antennas and Propagation
Volume 2018 (2018), Article ID 3949675, 9 pages
https://doi.org/10.1155/2018/3949675

Research Article

Codesigned Wideband High-Efficiency Filtering SIW Slot Antenna with High Selectivity and Flat Gain Response

Peng Kai Li,¹ Chang Jiang You,¹ Hong Fang Yu,¹ Yu Jian Cheng,² Yuan Wang Yang,¹ and Jian Hua Deng³

¹School of Information and Communication Engineering, University of Electronic Science and Technology of China, Chengdu 611731, China
²School of Electronic Science and Engineering, University of Electronic Science and Technology of China, Chengdu 611731, China
³School of Information and Software Engineering, University of Electronic Science and Technology of China, Chengdu 611731, China

Correspondence should be addressed to Chang Jiang You; cjyou@uestc.edu.cn

Received 26 December 2017; Revised 1 March 2018; Accepted 15 April 2018; Published 2 July 2018

Academic Editor: Giorgio Montisci

Copyright © 2018 Peng Kai Li et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Abstract.

A substrate integrated waveguide (SIW) slot antenna with wideband bandpass filtering performance is proposed in this journal. It consists of a SIW cavity, a transverse slot, three metal posts, and a SMA connector. Three metal posts split the SIW cavity into two TE₁₁₀-mode resonators. A long nonresonant transverse slot is utilized to realize a half TE₁₁₀-mode resonator and generate radiation simultaneously so as to reduce the size and release the use of extra radiator. Three in-band resonance poles and a radiation null at both out-of-bands are obtained. A prototype is fabricated and measured. Measured results demonstrate the proposed antenna is with a center frequency of 4.26 GHz, a fractional bandwidth of 9.1%, a high efficiency of 93%, and flat gain response and good skirt selectivity of 270 dB/GHz and 330 dB/GHz for the lower and upper out-of-band, respectively.

1. Introduction

Modern wireless communications demand the RF frontend system to be compact, lightweight, low cost, high efficiency, and multifunctional. In most of the RF front ends, the bandpass filter and the antenna are the key components whose performance will immediately affect the system performance. In the traditional systems, they are usually designed separately and connected by a 50 Ω or 75 Ω transmission line, which not only increases the volume but also maybe degrades in-band performance due to the mismatch and extra insertion loss caused by the interconnections [1]. Recently, a concept of filtering antenna has been proposed by integrating the bandpass filter and the antenna into a single component with filtering and radiating functions simultaneously. The proper integration of the antenna and filter turns out to be an efficient way to reduce the loss and enhance the efficiency of this functional block for a frontend system [2, 3].

Substrate integrated waveguide (SIW) technology has been effectively applied to design high-performance filters and antennas due to its low insertion loss [4–6]. Some filtering antennas based on SIW technology have been reported in [7–12]. In [7, 8], the antenna is planar and designed by cascading resonators and radiator. In [9–12], 3D configurations by placing the resonators under the radiator are applied. However, in reported works, they suffer from more insertion loss as they all need an extra radiator to cascade the filtering circuit and the filtering circuit is realized by full mode resonators which will lead to a large lossy circuit size. What is more, as they have no radiation null (transmission zero) in gain response or the radiation nulls are far away from the pass-band, the out-of-band skirt selectivity in the reported works is low.

To overcome these problems, a codesigned wideband filtering substrate integrated waveguide (SIW) slot antenna with high selectivity, high efficiency, and flat gain response is proposed in this journal. A long nonresonant transverse slot is utilized to realize a half mode resonator and generate radiation simultaneously so as to reduce the size and release the use of extra radiator. Three resonance poles are achieved to enhance the bandwidth and gain flatness. Two radiation nulls are introduced to increase the out-of-band skirt selectivity. Antenna work mechanism is explained, and the results and discussion are given.

2. Antenna Configuration and Mechanism

2.1. Configuration of the Proposed Antenna

Figure 1 shows the configuration of the proposed filtering antenna. The antenna is designed on a F4B-2 substrate with a thickness of 6 mm, a relative permittivity of 2.485, and a loss tangent of 0.0018. It is composed of a SIW cavity with size , a transverse slot with size etched on the top metal layer, three metal posts in the cavity and a SMA connector as feed. The metal posts are with a diameter of and space of . The antenna configuration is symmetrical about the y-axis. The antenna is designed and optimized at the centre frequency of 4.25 GHz. The detailed dimensions are as follows: , , , , , , , , , , and (unit: mm).

(a)

(b)

Figure 1: Configuration of the proposed filtering antenna. (a) Top view. (b) Side view.

The design evolution of the proposed antenna is illustrated in Figure 2. Firstly, a transverse slot is etched on the upper metal layer of SIW with end shorted (Ant. 1). Then, three metal posts are introduced at the right side of the slot (Ant. 2). Finally, a SIW cavity is formed and a SMA connector is used as feed (Ant. 3). It should be mentioned that the slot is nonresonant at the work band and its length is more than half a guide wavelength. The reflection coefficient S₁₁ and peak realized gain of Ant. 1–3 are given in Figure 3. From Figure 3(a), it can be found that the proposed antenna Ant. 3 has three resonance poles , , and . From Figure 3(b), it can be found that Ant. 1 has only one radiation null in gain response while Ant. 2 and Ant. 3 have two radiation nulls at almost the same frequencies. These phenomena will be explained in detail in the following section.

(a)

(b)

(c)

Figure 2: Design evolution. (a) SIW transverse slot. (b) SIW transverse slot with metal posts. (c) SIW cavity-backed transverse slot with metal posts.

(a)

(b)

Figure 3: Simulated results of Ant. 1, Ant. 2, and Ant. 3. (a) Reflection coefficient . (b) Peak realized gain.

2.2. Antenna Mechanism of Radiation Nulls

The configuration of a SIW transmission line with end shorted is shown in Figure 4(a). When it is excited by TE₁₀ mode, the surface current is with standing wave distributions, as expressed by

(a)

(b)

Figure 4: SIW transmission line with end shorted and surface current distributions. (a) Configuration of SIW transmission line with end shorted. (b) Surface current distributions at different frequencies.

Here, is defined as the maximum current magnitude, denotes the phase constant at frequency , and means the distance from the shorted end. Figure 4(b) gives the surface current distribution at two different frequency and, which are labelled as case 1 and case 2, respectively. If a slot is etched on the top layer, it will generate radiation or not which depends on the current distribution at the position of the slot. It should be stated that there will be no radiation at the current null. Thus, if the slot is located at the position of current null as case 1 in Figure 4(b), there will be no radiation generated from the slot. And if the slot is not located at the position of current null as case 2, there will be radiation generated from the slot. Hence, a radiation null will occur at the frequency where the current at the slot satisfies the following condition that

That is the reason why one radiation null is generated in Ant. 1, which is also explained in [13].

Actually in Ant. 2 and Ant. 3, when three metal posts are introduced, a resonator is formed. The configuration of SIW transmission line loaded by a resonator and its equivalent circuit model is illustrated in Figure 5(a). The resonator can be simply modelled as shunt inductor and capacitor, and it resonates at 4.25 GHz. The phase shift at the reference plane in the one end shorted structure is twice that of a two-port resonator model as discussed in [14]. Thus, as shown in Figure 5(b), the phase shift of the end shorted resonator can be expressed as

(a)

(b)

Figure 5: Illustration of analysis model and the results. (a) Analysis model of an end shorted resonator. (b) Simulated phase of .

For Ant. 2 and Ant. 3, the resonator can be considered as the load of the transmission line, as illustrated in Figure 6(a). The phase difference at the reference plane can be written as

(a)

(b)

Figure 6: Illustration of analysis model and the results. (a) Analysis model of Ant. 2. (b) Simulated phase difference.

Here, denotes the distance from the three metal posts. According to transmission line theory [15], the current strength at the reference plane can be expressed as

Here, means the voltage of incident wave, denotes the characteristic impedance of the transmission line, and is voltage reflection coefficient. According to (5), the minimum current strength (current null) satisfies the condition that

According to (3), (4), and (6), when current null occurs, the minimum should meet that

It means when is about ( is guide wavelength), two current nulls will occur at both sides of the resonance. Figure 6(b) gives the phase difference when changes near (13 mm). The phase difference is modularized by 2π and is limited to the [−180° and 180°] range. It can be found when , there exist two frequency and at both sides of 4.25 GHz. From the abovementioned analysis, the frequency where is the frequency of current null. It can also be seen from Figure 6(b), frequencies of the two current nulls shift downward when increases from 11 mm to 15 mm. It means the upper current null will come close to the resonance, and the lower current null will be away from the resonance.

According to transmission line theory, input impedance is defined as . When the current is minimum, the input impedance is maximum. So the current null can be deduced from the feature of input impedance. Figure 7 gives the simulated magnitude of input impedance with the variation of . It can be found that one impedance peak is observed at both sides of 4.25 GHz. The frequency of the impedance peak is the frequency of current null. Due to the lossy material used in the simulation, when approaching the resonance, the loss is more and voltage reflection coefficient becomes smaller. Meanwhile, the minimum current strength becomes larger according to (5). It means the impedance peak becomes smaller near the resonance 4.25 GHz as can be seen in Figure 7. So the depth of the two radiation nulls in Ant. 2 and Ant. 3 can be estimated from Figure 7. A larger impedance peak means a deeper radiation null, as can be proved from the simulated peak realized gain of Ant. 3 with different in Figure 8.

Figure 7: Simulated magnitude of illustrated in Figure 5(a).

Figure 8: Simulated peak realized gain of Ant. 3 with different .

To achieve two symmetrical radiation nulls about 4.25 GHz, here in the proposed antenna, the slot is located at the position where is chosen to be 13 mm which is about a quarter phase wavelength at the resonant frequency of .

2.3. Antenna Mechanism of Resonance Poles

As can be seen in Figure 3(a), three resonance poles , , and are generated in Ant. 3. Actually, when three metal posts are introduced, the large cavity is divided into two TE₁₁₀-mode resonators. What’s more, when a long slot is cut, a half TE₁₁₀-mode resonator is created. The three resonant poles are generated by the couplings between the three resonators, and they can be explained by the basic resonant mode superposition. Figure 9 gives the simulated H-filed at different phase of the three resonance poles and their superposition modes. The modes created by the two TE₁₁₀-mode resonators and their coupling can be considered as an even mode and an odd mode [16]. Thus, as illustrated in Figure 9(a), the first resonance pole can be considered as the superposition of a half TE₁₁₀ mode and an even mode. As illustrated in Figure 9(b), the second resonance pole can be considered as the superposition of a half TE₁₁₀ mode and a TE₁₁₀ mode. As illustrated in Figure 9(c), the third resonance pole can be considered as the superposition of a half TE₁₁₀ mode and an odd mode. It should be noticed that all the resonance poles are in relation to the half TE₁₁₀ mode. This is because only the half TE₁₁₀ mode can generate radiation through the slot, while at even and odd mode, the slot is almost located at the position of current null. It is innovative that in the proposed design, the slot not only acts as a radiator but also is essential to the formation of the half mode resonator. This merged structure is different to the commonly used method which will introduce extra radiator.

(a)

(b)

(c)

Figure 9: Illustration of the mechanism of resonance poles in Ant. 3. (a) First resonance pole . (b) Second resonance pole . (c) Third resonance pole .

3. Results and Discussion

The proposed antenna is fabricated and measured, and the photograph is shown in Figure 10. The simulated and measured reflection coefficient , realized gains, and total efficiencies are illustrated in Figure 11. It can be found that measured bandwidth () is 9.1% (4.09–4.48 GHz), agreeing well with the simulated one of 10.1% (4.05–4.48 GHz).

Figure 10: Photograph of the fabricated antenna.

Figure 11: Simulated (dashed lines) and measured (solid lines) reflection coefficient , realized gains, and total efficiencies.

The measured and simulated average realized gains over the work band are 6.2 dBi and 6.3 dBi, respectively. The gain response is flat over the work band with ripples less than 0.4 dB, which is attributed to the nonresonant slot radiation. The bandpass filtering performance is remarkable, as the efficiency over the work band reaches to a maximum as high as 93% which means insertion loss of about only 0.3 dB is introduced, while the out-of-band efficiency approaches zero. The high efficiency is benefited from the nonresonant slot radiation and small lossy circuit area as a half mode resonator is realized and no extra radiator is utilized. As can be observed in Figure 11, two measured radiation nulls at 4.04 GHz and 4.56 GHz in gain response are very close to the pass-band, so high skirt selectivity is achieved. The measured lower and upper out-of-band skirt selectivity is of 270 dB/GHz and 330 dB/GHz, respectively.

The simulated and measured radiation patterns at the three resonance poles 4.12 GHz, 4.26 GHz, and 4.46 GHz are shown in Figure 12. It can be seen almost symmetrical radiation patterns are observed in E/H plane. The slightly tilts at E-plane patterns are mainly caused by the off-centered slot position. Good front-to-back ratios and cross polarization levels of more than 20 dB are obtained at both planes. The simulated copolarization patterns are in accordance with the measured ones, and the copolarization patterns are stable over the work band. The discrepancy in cross polarization between simulated and measured value is mainly attributed to the slightly tilted antenna placement during measurement.

(a)

(b)

(c)

Figure 12: Simulated and measured radiation patterns at three resonance poles. (a) 4.12 GHz. (b) 4.26 GHz. (c) 4.46 GHz.

A comprehensive comparison with previous works utilizing SIW slot is summarized in Table 1. It can be found in our work the advantages of wideband, high selectivity, and high efficiency (Eff.) compared to other works. The high efficiency can also be deduced from the lossy circuit area which can be evaluated by size × layer. It is obvious that our work has the least value of size × layer.

Table 1: Comparisons with previous reported works.

Ref
Number of Pole
Number of Null
BW (%)
Selectivity (dB/GHz)
Eff. (%)
Size ()
Layer

[8]
4
0
6
113/74
89
1.3 × 1.7
1

[9]
2
0
3
63/28
58.9
0.9 × 0.9
2

[10]
3
2
1.6
96/55
86
0.7 × 0.9
3

[11]
3
0
5.5
26/57
N.A.
1.4 × 1.6
2

Ours
3
2
9.1
270/330
93
1.1 × 1.2
1

is the guide wavelength at the centre frequency of the work band. Selectivity is calculated as , where is the attenuation of the first out-of-band radiation null or 20 dB attenuation if no radiation null and is the 3 dB attenuation; and are their corresponding frequency.

4. Conclusion

A substrate integrated waveguide slot antenna with bandpass filtering performance in gain response is presented in this journal. A long nonresonant slot is introduced to realize a half mode resonator and generate radiation so as to reduce the size and release the use of extra radiator. Two radiation nulls are generated to enhance the selectivity. The measured results show that wide bandwidth of 9.1%, high selectivity of 270/330 dB/GHz for lower/upper out-of-band, and high efficiency of 93% and flat gain response are obtained in the proposed antenna. The high performance indicates it is a good candidate for the functional module integrated with filter and antenna in RF front system.

Conflicts of Interest

The authors declare that they have no competing interests.

Acknowledgments

This work was supported by Fundamental Research Funds for the Central Universities under Grant ZYGX2016J006.

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