A Low Cross-Polarization Microstrip Antenna Array for Millimeter Wave Applications

. A low cross-polarization microstrip antenna array for millimeter wave (mmW) applications is proposed in this paper. Te antenna element is composed of symmetric T-shaped patches with vias. Te adoption of a double-sided symmetric radiation patch structure can suppress the cross-polarized electric feld, and the vias reduces energy leakage during the power transmission along the antenna patches. To verify the concept, a 1 × 8 antenna array is fabricated and measured. Te measured − 10dB impedance of the antenna array is 28.4% (31GHz–41.5 GHz) and the peak gain is 15 dBi. Te cross polarization ratio is above 35dB and the 3dB beamwidths on E -plane and H -plane are 7.5 ° and 135 ° , respectively. Te proposed compact size and low cross-polarization antenna array might be a good choice for phased array radar and 5th generation (5G) mobile communication applications.


Introduction
In recent years, the millimeter wave (mmW) band has developed vigorously in related technologies and applications due to its rich spectrum resources, especially in 5G big data transmission, radar, and imaging applications. In practical applications, the mmW communication systems pursue easy integration and high reliability. Hence, the antenna designs for mmW applications are also required to possess the characteristics of the compact structure and easy integration.
Te mmW antennas using cavity or waveguide structures have been extensively studied due to their low loss [1][2][3]. However, the big dimensions and weights restrict its applications for large-scale arrays' integration. In contrast, PCB antennas made of low loss dielectric are a good choice for millimeter front-end systems. Many researchers have realized the broadband dual polarization/circular polarization characteristics utilizing multilayer stacking multiple resonance branches/gaps [4][5][6][7][8][9][10][11][12]. A broadband magnetic electric (ME) dipole antenna with circular polarization operating in the mmW band is proposed in [12]. Te rectangular patch is fed through the cross gap of the ground plane to realize the circular polarization. A ME dipole antenna consisting of two patches and a short strip is applied for Ka band [13]. Te patch is fed by a rectangular slot on the lower layer of the substrate-integrated waveguide (SIW). Te 4 × 4 array is designed and achieves 5.4 GHz bandwidth (27.7-32.3 GHz) and 15.25 dB gain. Tere are also studies on single-layer microstrip or SIW type antennas [14][15][16][17][18][19][20][21][22][23][24][25]. An improved bow tie symmetric dipole linear array antenna with single-layer plate structure is proposed in [25]. Bandwidth of 10.51 GHz (23.41-33.92 GHz) and 10.7 dBi high gain is achieved.
In this paper, a mmW linear array antenna with ground refection is proposed, and its structure consists of strip lines and double-sided patches. Te radiation element consists of a symmetrical T-shaped patch and vias. Te antenna is fed by η-type microstrip line. Te 1 × 8 linear array antenna is then designed. Te proposed antenna achieves the bandwidth of 10.5 GHz (31 GHz-41.5 GHz, 28.4%) with above 35 dB cross-polarization ratio performance. Te 3 dB beamwidth is 7.5°on E-plane and the realized gain of the antenna array is over 14.5 dB. Moreover, 135°wide beam characteristic on the H plane are also obtained. Terefore, the proposed antenna array possesses a compact structure and would be an excellent candidate for phased array radar and 5G mobile communication applications.

Antenna Element Design
In this paper, the antenna element is designed using Rogers RT Duroid 5880 printed circuit board (PCB). Te thickness of the board is 0.254 mm and the dielectric constant is 2.2. Te antenna radiation element adopts a double-sided symmetrical T-shaped structure with a rectangular groove, as shown in Figure 1. An η-type microstrip in the middle of the two PCB layers is introduced acting as the coupling fed structure. A refective ground plane is introduced at the bottom of the antenna element to suppress the back radiation, thus reducing the antenna back lobe. Te distance from the top of the antenna to the refector s_h2 is usually a quarter of the wavelength, which increases the realized gain of the antenna element by making the refective electric feld cophase stacking, as shown in Figure 2. Te vias along the dipoles is proposed to reduce the energy difusion within the substrates. Te energy difusion in the inefective area of the microstrips is signifcantly reduced, as shown in Figure 3. Te reduction of inefective energy difusion is benefcial to reduce the coupling efect between the antenna inner feeding network and other circuits within the overall system, especially in the millimeter wave band. Te doublesided symmetrical structure can efectively eliminate the cross-polarization electric feld and therefore improves the cross-polarization performance of the antenna. It is because the cross-polarization displacement current generated between the feed microstrip and double-layer patches in the double-sided symmetrical structure is ofset by the equal magnitude and the opposite vector direction, while the single-layer PCB board structure is not. Te cross-polarized electric feld distributions of single-layer and double-layer radiation patches are shown in Figure 4. It is obvious that the electric feld energy at the radiation interface of double-layer patches is signifcantly smaller. Also, the cross-polarization of double-layer patches is about 20 dB lower than the singlelayer patch, as shown in Figure 5. Trough optimization, the parameters of the antenna element have to be optimized and are listed in Table 1.
Te length scanning result of the parameter f_h is shown in Figure 7. Te parameter f_h mainly afects the impedance bandwidth matching of the antenna and the resonant frequency. According to the optimization results, the parameter f_h is selected as 1.64 mm. Te antenna element achieves a bandwidth of 12.5 GHz, and the impedance bandwidth ratio is 33.8% at 37 GHz.
Te surface current distributions of diferent resonant frequencies are shown in Figure 8. It can be found that the vias avoids the difusion of the energy inside the plate. Te vias can also provide an extended path for the T-shaped patch. Te copolarization and cross-polarization patterns of the antenna element at 37 GHz are shown in Figure 9. Te antenna gain reaches 6.83 dB, and the 3 dB beamwidth on Eplane is 68°, the beamwidth on H-plane is 135°, and the cross-polarization ratio is 40 dB.
On one hand, the refector plane refects back radiation energy and superposes it with the forward propagated energy. Teoretically, the distance is 1/4 wavelength of the central frequency, which achieves the in-phase superposition. On the other hand, the antenna gain is the largest, while the 3 dB beam width is the smallest. At 37 GHz, the 1/4 wavelength is 2 mm. Comparison of H-plane pattern versus s_h2 (varying from 2 mm to 4 mm) is shown in Figure 10, where s_h2 is the distance from the antenna aperture to the refector plane. When s_h2 is 2 mm, the maximum gain of the antenna element is 8.2 dB, while the 3 dB beamwidth is the minimum 91.5°. Te gain decreases as s_h2 becomes larger, while the beam width is broadened. According to the optimization process, s_h2 is set as 3 mm and c_w is set as 3.5 mm.

Antenna Array Design
A mmW linear array antenna with dual side dipoles of the microstrip structure is shown in Figure 11. A 1 × 8 linear array antenna is designed. Te distance between the nearby elements is 7 mm. Structure of the feed network is shown in Figure 12. Te energy difusion in the inefective area of the   E Field (V_per_m feed network is signifcantly reduced by adding metallic vias, as shown in Figure 13, and the transmission coefcient of the feed network with metallic vias is shown in Figure 14. Te overall dimension of the antenna array is only 58 mm × 9.3 mm if the transition structure from microstrip to coaxial line is not considered. Snap of the proposed antenna array is shown in Figure 15. Te microstrip feeding port can be easily integrated with the transmitter and receiver (T-R) component.

Antenna Simulation and Measurement
Te simulation and measurement results of VSWR for the antenna array are shown in Figure 16. Te antenna array achieves the bandwidth of 10.5 GHz (31 GHz-41.5 GHz, VSWR ≤ 2). Figure 17 shows the simulated and measured patterns of the antenna at 31 GHz, 37 GHz, and 41 GHz. It can be seen that the simulated results are compatible with the measured results. Te proposed antenna has above 35 dB cross-   Note that the parameter s_w afects the resonant frequency of the antenna. Te parameter p_h is the length of η-type feeding microstrip and f_h is the length of the rectangular groove. Te parameter p_h mainly afects the resonant frequency of the antenna. In order to obtain the 37 GHz resonant frequency and meet the bandwidth characteristics requirement, p_h � 1 mm is set. Te S 11 versus parameter p_h is shown in Figure 6. Unit: mm.   polarization performance in the whole operating frequency band. Te E-plane pattern has a 3 dB beamwidth of 7.5°an a side lobe level of −13 dB, while the H-plane pattern has a wide beam width of 135°. Figure 18 shows the simulation and measurement results of antenna gain. Te proposed antenna has a high gain of 15 dB.     Figure 11).

Conclusions
In this paper, a T-shaped linear array antenna with refective ground operating at the mmW band is proposed. Protype measurement of the antenna shows that in the bandwidth of 10.5 GHz (31 GHz-41.5 GHz), stable high gain and good radiation pattern characteristics can be obtained. Moreover, the antenna realized a compact size with low cross-polarization. Te proposed antenna is suitable for mmW band-phased array applications and 5G communication.

Data Availability
All data, models, and code generated or used during the study are included within the article.

Conflicts of Interest
Te authors declare that they have no conficts of interest.